Direct conversion receiver having a low pass pole implemented with an active low pass filter

ABSTRACT

A wireless communications mobile station ( 10 ) includes at least one antenna ( 240 ) and a RF transceiver ( 210,220 ) containing a direct conversion receiver ( 1 ) coupled to the antenna. The direct conversion receiver contains a low noise amplifier ( 3 ) for amplifying a received RF signal and for outputting the amplified RF signal to a current switching down-conversion mixer ( 4 ). The down-conversion mixer has a first input node for receiving the amplified RF signal, a second input node for receiving a local oscillator (LO) signal for mixing with the amplified RF signal and an output node coupled to an input of an operational amplifier forming a low pass filter ( 5 A). In accordance with an aspect of this invention the low pass filter has a low pass pole generated by a resistor R and a capacitor C coupled in parallel in a feedback path of the operational amplifier, where a low pass comer frequency of the low pass filter is inversely proportional to the product of R and C. In a preferred embodiment at least the down-conversion mixer and the low pass filter are implemented as part of an integrated circuit, and the resistor and the capacitor are fabricated within the integrated circuit.

TECHNICAL FIELD

The teachings of this invention relate generally to radio frequency (RF)receivers and, more specifically, relate to direct conversion RFreceivers and to mixers used with direct conversion RF receivers.

BACKGROUND

Direct conversion RF receivers, or more simply direct conversionreceivers (DCRs), have been found to be attractive for use in massproduced consumer communication products such as cellular telephones andpersonal communicators. This is due to the fact that the DCR has arelatively simple structure (as compared to the conventional heterodyneor superheterodyne type of receiver), a reduced component count, andenables a high level of circuit integration to be achieved. The DCR isso-named because the received RF frequency is down-mixed ordown-converted directly to a DC baseband signal, or to a very lowfrequency baseband signal, without undergoing one or more stages ofintermediate frequency (IF) down-mixing.

FIG. 1 is a simplified schematic diagram that illustrates a prior artexample of a DCR 1 used for receiving a modulated RF input signal anddownconverting it to in-phase (I) and quadrature (Q) baseband signals.The DCR 1 includes an input RF passband filter 2, a low noise amplifier(LNA) 3 and I and Q channels or branches each containing adown-conversion mixer 4, a low pass filter (LPF) 5, an automatic gaincontrol (AGC) block 6, a second LPF 7 and an analog to digital converter(ADC) 8. The mixing frequency input to the I and Q mixers is supplied bya local oscillator (LO) 9 containing a phase locked loop (PLL) 9A, avoltage controlled oscillator (VCO) 9B, a buffer amplifier 9C andpossibly a divider 9D. The divider 9D may also conveniently provide thedesired ninety degree phase shift between the LO signals input to the Iand Q branch mixers 4. The LO signal is typically equal to (or very nearto) the center frequency of the RF input signal (note that theinstantaneous frequency of the RF input signal can differ from the LOfrequency due to modulation).

It is important to low pass filter the down-converted signal as soon aspossible after the mixers 4, otherwise the signal will exhibit linearityand compression-related problems upon further amplification. This is dueto adjacent channel signal that may be at a higher level than thedesired channel (own channel) signal. The use of low pass filteringserves to attenuate the adjacent channel signals, thereby allowing foradditional amplification of the own channel signal. This type ofoperation is typically referred to in the art as channel filtering.

FIG. 2A is a simplified schematic diagram of a prior art embodiment ofthe mixer 4 and first LPF 5 of FIG. 1 (a Gilbert Cell mixer). For thepurposes of this discussion it does not matter whether the I branch orthe Q branch circuitry is illustrated. Note that differential signalsand circuitry are illustrated and are assumed to be used. Thedifferential RF input signal from the LNA 3 is applied to a pair oftransconductors 4A and 4B. The outputs of transconductors 4A and 4B arecoupled to a switch pair (SW1, SW2) that also receives the LO signal.Common mode DC and input signal dependent differential currents,produced by the transconductors 4A and 4B, are alternated to the outputloads according to the LO signal, thereby causing downconversion. Thecurrents are converted to voltages in resistors R1 and R3 (R1 istypically equal to R3). However, conversion results at a high frequencyare attenuated in the low pass filter formed by R1, R3 and C1 (LPF 5).The output is connected to the next stage, shown as the AGC block 6 inFIG. 1.

In the illustrated prior art design for the LPF 5 the low pass cornerfrequency is given by 1/(2π(2(R1*C1)), assuming that R1=R3. However, theohmic value of R1 and R3 cannot be made large as an undesirable large DCvoltage drop will result. This is true because R1 and R3 are in serieswith the mixer 4 and the DC power supply, and the mixer DC current(I_(DC)) flows through R1 and R3. The conventional solution insteadincreases the value of C1. This solution, however, creates a problemwhen it is desired to fabricate the DCR in an integrated circuit form,as the required large value of capacitance for C1 correspondinglyrequires a substantial amount of integrated circuit area to implement.An alternative solution is to make C1 an external discrete component,but this approach requires that additional integrated circuit pins beprovided, and thus adds cost, increases fabrication/testing complexity,and introduces a possible interference source and imbalance due topin-related parasitics.

FIG. 2B shows a conventional Low Pass Filter (LPF) 5′ constructed usingan active component, i.e., an operational amplifier (op amp), while FIG.2C shows the roll-off in gain as a function of frequency. Note that theinput (voltage mode) is applied through R1 and R3.

Reference may be had to a publication entitled: “A 1.5 GHz Highly LinearCMOS down conversion mixer”, IEEE Journal of Solid State Circuits, J.Crols et al., Vol. 30, No. 7, July 1995. This publication describes aCMOS mixer topology that uses two additional capacitors added to theconventional CMOS lowpass filter structure, enabling GHZ signals to beprocessed while using a low frequency operational amplifier (op amp) asan output amplifier.

Reference can also be made to B. Song, “CMOS RF Circuits for DataCommunications Applications”, IEEE Journal of Solid-State Circuits, Vol.SC-21, No.: 2, April 1986, pps. 310–317, for showing a triode low passmixer.

SUMMARY OF THE INVENTION

The foregoing and other problems are overcome by methods and apparatusin accordance with embodiments of this invention.

A wireless communications mobile station includes at least one antennaand a RF transceiver containing a direct conversion receiver coupled tothe antenna. The direct conversion receiver contains a low noiseamplifier for amplifying a received RF signal and for outputting theamplified RF signal to a down-conversion current-switching mixer, whichcould be implemented as a bipolar mixer. The down-conversion mixer has afirst input node for receiving the amplified RF signal, a second inputnode for receiving a local oscillator signal for mixing with theamplified RF signal and an output node coupled to virtual ground of anoperational amplifier, which forms a low pass filter.

In accordance with an aspect of this invention the operational amplifierhas a low pass pole generated by a resistor R and a capacitor C coupledin parallel in a feedback path of the operational amplifier, where a lowpass corner frequency of the low pass filter is inversely proportionalto the product of R and C.

In a preferred embodiment at least the down-conversion mixer and the lowpass filter are implemented as part of an integrated circuit, and theresistors and the capacitors are fabricated within the integratedcircuit, thereby reducing component count and cost while increasingreliability.

In a preferred embodiment the direct conversion receiver has an in-phase(I) branch and a quadrature (Q) branch following the low noiseamplifier, and each of the I and Q branches are constructed to containone of the down-conversion mixers and one of the low pass filters.

A method of operating a direct conversion receiver integrated circuit inaccordance with this invention includes amplifying an input RF signal;down-converting the amplified RF signal to a down-converted signal usinga mixer having an input transconductor stage for receiving the amplifiedRF signal and a mixer core. The mixer processes the RF signal in thecurrent mode and processes a local oscillator signal in the voltagemode, and the method low pass filters the down-converted signal with anactive low pass filter having a low pass pole generated by an amplifierhaving a resistor R and a capacitor C coupled in parallel in a feedbackpath of the amplifier. A low pass corner frequency of the low passfilter is inversely proportional to the product of R and C.

In a still further aspect this invention provides a receiver circuit foruse in a mobile station, such as a cellular telephone or a personalcommunicator. The receiver circuit includes a Gilbert Cell mixer thatreceives an RF signal to be mixed with a local oscillator signal. TheGilbert Cell mixer has first and second outputs connected to first andsecond inputs, respectively, of an operational amplifier for providinginput signal dependent differential currents i_(diff) to the first andsecond inputs of the operational amplifier. The first and second inputsof the operational amplifier are connected to a supply voltage throughfirst and second resistances, respectively, and receive common modecurrents I_(DC) therefrom. The operational amplifier has a firstparallel RC network coupled between a first output node of theoperational amplifier and the first input, and has a second parallel RCnetwork coupled between a second output node of the operationalamplifier and the second input. The operational amplifier and the firstand second parallel RC networks form a low pass filter at the first andsecond outputs of the Gilbert Cell mixer. The first and second outputsof the Gilbert Cell mixer are connected to virtual ground nodes of theoperational amplifier.

BRIEF DESCRIPTION OF THE DRAWINGS

The above set forth and other features of these teachings are made moreapparent in the ensuing Detailed Description of the PreferredEmbodiments when read in conjunction with the attached Drawings,wherein:

FIG. 1 is a simplified schematic diagram that illustrates a prior artexample of a direct conversion receiver;

FIG. 2A is a simplified schematic diagram of a prior art embodiment ofthe mixer and low pass filter shown in FIG. 1;

FIG. 2B shows a conventional Low Pass Filter (LPF), while FIG. 2C showsthe roll-off in gain as a function of frequency;

FIG. 3 is a simplified schematic diagram of an embodiment of the mixerand low pass filter in accordance with the teachings of this invention;

FIG. 4 is a more detailed schematic diagram of the embodiment of themixer and low pass filter in accordance with the embodiment shown inFIG. 3; and

FIG. 5 is a block diagram of an embodiment of a wireless communicationssystem that includes a mobile station having the improved directconversion receiver illustrated in FIGS. 3 and 4.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Reference is now made to FIG. 3 for showing a simplified schematicdiagram of an embodiment of the mixer 5 and an improved low pass filter5A in accordance with the teachings of this invention. The outputcurrents from the mixer core 4 are connected to the virtual ground nodeof the operational amplifier (op amp) 5B. Because of feedback bothinputs to the op amp 5B are at the same DC potential, and differentialcurrents are forced to go through the feedback circuitry. However,common mode DC currents can also be supplied by R1 and R3. This ispreferred, as it means that the comer frequency of the low pass pole isnot affected by the values of R1 and R3. As such, their values can bemade small, thereby producing a low DC voltage drop from the supplyvoltage VCC.

In this embodiment the low pass pole of the LPF is implemented not bythe passive LP filter comprised of R1, R3 and C1, but instead by anactive LP filter comprised of the operational amplifier 5B incombination with the paralleled feedback components R2/C2 and R4/C3. Thevalues of R2 and R4 are thus not limited by the DC voltage drop causedby DC currents, and they can therefore be made larger in value.Consequently, the value of C2 and C3 can be less than the capacitancevalue of C1.

Note that in this embodiment, and contrasting same with the prior artshown in FIG. 2B, the normal input port (left sides of R1 and R3) isconnected to VCC (thereby providing common mode current I_(DC)), and theinput signal (current mode i_(diff)) is applied to the inputs of theoperational amplifier 5B.

Note as well that this embodiment enables to the values of R1/R3 andR2/R4 to be optimized.

FIG. 4 is a more detailed schematic diagram of the embodiment of themixer 4 and the improved first LPF 5A in accordance with the embodimentshown in FIG. 3, and shows in greater detail the construction of themixer 4 and its coupling to the LPF 5A. The mixer 4 is based on bipolartransistors, having in RF input transconductors formed by transistorsT1,T2 and emitter resistors R10,R11, and a mixer 4 core comprised oftransistors T3, T4, T5 and T6. The differential outputs from the LNA 3(RF+, RF−) are coupled to the bases of NPN transistors T1 and T2,respectively, which are connected through their emitters to groundthrough resistances R10 and R11, respectively. The details of thebiasing of T1 and T2 are not shown to simplify the drawing. Thecollectors of T1 and T2 are connected to the emitters of differentialtransistor pairs T3, T4 and T5, T6, respectively, having their baseterminals connected to the differential LO input signals LO+ and LO− asshown. Load resistances R1 and R3 are connected between the collectorsof T3, T5 and T4, T6, respectively, and the positive voltage supply VCC.The inputs to the op amp 5B are connected between R1 and R3 and thecollectors of T3, T5 and T4, T6, respectively. An additionalcapacitance, C4, is connected between the inputs to op amp 5B to serveas a filter for removing high frequency mixing components. The value ofC4 can be small, as filtering is provided for only very highfrequencies. C4 may also be implemented as two capacitances, shown indashed outline as C4A and C4B, placed in parallel with R1 and R3,respectively. The combination of C4, C4A and C4B can be used together ifdesired, or C4 may be eliminated. If present, the combination of C4 andthe op amp 5B functions beneficially to attenuate higher frequencysignals at the output of the mixer 4, as the op amp 5B has a finitefrequency response and may not itself adequately attenuate the higherfrequency signals. In any case, the amplified and filtered mixer outputsignal is provided at the output of op amp 5B.

In the improved active LPF 5A shown in FIGS. 3 and 4 the low pass cornerfrequency is inversely proportional to the product of R2 and C2,assuming that R4=R2 and C3=C2. However, since R2 is not in series withthe positive voltage supply, as is R1 in FIG. 1, it can be made large invalue, enabling the value of C2 to be made small, thereby conservingintegrated circuit area when fabricating C2 on-chip.

Representative and non-limiting values for the components shown in FIGS.3 and 4 are as follows: R1=R3=500 ohms, C4=20 pF, R2=R4=20 kohms andC2=C3=50 pF. By contrast, representative component values for the priorart circuit solution shown in FIG. 2A are as follows: R1=R2=500 ohms,and C1=1 nF. The significant reduction in the total capacitance value,120 pf vs. 1000 pf, results in the realization of the benefits discussedabove, such as the decrease in required circuit area to implement theDCR 1 in an integrated circuit embodiment. Relatedly, since thecapacitance values are in the range of tens of picofarads, thecapacitors can be implemented on-chip, and additional pins are notrequired to be provided for connecting to an external discretecapacitance. Note as well that the operation of the mixer 4 and the LPFis improved through the use all integrated components, as bettermatching and reproducibility is achieved as compared to the use ofdiscrete components, and parasitics related to the presence of externalintegrated circuit pins can be eliminated. In addition, the signal levelat the mixer 4 output (input to op amp 5B) is attenuated, enabling areduction in the power supply voltage and/or a more linear signal rangeat the input to the mixer 4.

Reference was previously made to the publication entitled: “A 1.5 GHzHighly Linear CMOS down conversion mixer”, IEEE Journal of Solid StateCircuits, J. Crols et al., Vol. 30, No. 7, July 1995, for describing aCMOS mixer topology that uses two additional capacitors added to theconventional CMOS lowpass filter structure and a low frequencyoperational amplifier as an output amplifier.

As can be appreciated in light of the foregoing description of thisinvention, the mixer of Crols et al. differs significantly from themixer 4, as the mixer of Crols et al. operates in the MOSFET trioderegion as opposed to the mixer 4 which employs a mixing core thatswitches the input currents introduced by the transconductance stage.That is, the prior art mixing transistors of Crols et al. operate withthe RF and LO signals both in the voltage mode, whereas the mixing coreof mixer 4 operates with the RF signal in the current mode and the LOsignal in voltage mode. The mixer of Crols et al. also does not employthe resistors R1 and R3, which provide a relatively high DC currentthrough the transistors of the mixer core (T3, T4, T5, T6) and thetransconductance stage (T1 and T2).

An example of the use of the improved DCR 1 is shown in FIG. 5, whichillustrates a simplified block diagram of an embodiment of a wirelesscommunications system that is suitable for practicing this invention.The wireless communications system includes at least one mobile station(MS) 100. FIG. 5 also shows an exemplary network operator 20 having, forexample, a network node 30 for connecting to a telecommunicationsnetwork, such as a Public Packet Data Network or PDN, at least one basestation controller (BSC) 40, and a plurality of base transceiverstations (BTS) 50 that transmit in a forward or downlink direction bothphysical and logical channels to the mobile station 100 in accordancewith a predetermined air interface standard. A reverse or uplinkcommunication path also exists from the mobile station 100 to thenetwork operator, which conveys mobile originated access requests andtraffic. Communications may occur in macrocells or in microcells,depending on the nature of the network operator 20. The air interfacestandard can conform to any suitable standard or protocol, and mayenable both voice and data traffic, such as data traffic enablingInternet 70 access and web page downloads.

The mobile station 100 typically includes a microcontrol unit (MCU) 120having an output coupled to an input of a display 140 and an inputcoupled to an output of a keyboard or keypad 160. The mobile station 100may be a handheld radiotelephone, such as a cellular telephone or apersonal communicator. The mobile station 100 could also be containedwithin a card or module that is connected during use to another device.For example, the mobile station 10 could be contained within a PCMCIA orsimilar type of card or module that is installed during use within aportable data processor, such as a laptop or notebook computer, or evena computer that is wearable by the user.

The MCU 120 is assumed to include or be coupled to some type of a memory130, including a read-only memory (ROM) for storing an operatingprogram, as well as a random access memory (RAM) for temporarily storingrequired data, scratchpad memory, received packet data, packet data tobe transmitted, and the like. The ROM may store a program enabling theMCU 120 to provide a suitable user interface (UT), via display 140 andkeypad 160, with a user. Although not shown, a microphone and speakermay be provided for enabling the user to conduct voice calls in aconventional manner.

The mobile station 100 also contains a wireless section that includes adigital signal processor (DSP) 180, or equivalent high speed processoror logic, as well as a wireless transceiver that includes a transmitter200 and a receiver 220, both of which are coupled to at least oneantenna 240 for communication with the network operator 20.

In the presently preferred embodiment the receiver 220 is constructed tocontain a DCR of a type generally shown in FIG. 1, but modified toinclude the improved LPF 5A shown in FIGS. 3 and 4. At least one localoscillator (LO) 9, as shown in FIG. 1, is provided for tuning thetransceiver, in particular the DCR 1. Data, such as digitized voice, andpacket data, is transmitted and received through the antenna 240.

While these teachings have been particularly shown and described withrespect to preferred embodiments thereof, it will be understood by thoseskilled in the art that changes in form and details may be made thereinwithout departing from the scope of this invention. For example, whiledescribed above in the context of a DCR that employs differentialsignals, single ended signal embodiments are also within the scope ofthese teachings. Also, these teachings can be employed in DCRembodiments that do not provide in-phase and quadrature channels, butonly a single channel. This invention can be employed in DCR embodimentsthat generate a zero Hz (DC) down-converted signal, as well as in DCRembodiments known as low IF architectures where the down-convertedsignal is not a DC signal (e.g., one having a frequency a few hundredHertz or more). Furthermore, this invention is not to be construed to belimited to the specific component values that were given above by way ofillustration. In addition, all of the components (resistors andcapacitors) can be made adjustable or trimmable in order to enable thecorner frequency to be adjusted to the required tolerances. Furthermore,this invention can be realized with a transconductance stage and aswitching mixer that are implemented with any of a number of devicetechnologies including, but not limited to, bipolar, MOSFET and MESFET.Thus, this invention should be construed as having a scope commensuratewith the scope of the appended claims, and equivalents thereof.

1. A direct conversion receiver comprising a down-conversion currentswitching mixer comprising a first input node for receiving an RFsignal, a second input node for receiving a local oscillator signal formixing with the RF signal and an output node coupled to an input of anamplifier forming a low pass filter having a low pass pole generated bya resistor R and a capacitor C coupled in parallel in a feedback path ofsaid amplifier, said output node further being coupled to a power supplyvoltage through load resistances.
 2. A direct conversion receiver as inclaim 1, where a low pass corner frequency of said low pass filter isinversely proportional to the product of R and C.
 3. A direct conversionreceiver as in claim 1, where said down-conversion mixer operates as adifferential signal down-conversion mixer having a pair of first inputnodes, a pair of second input nodes, and first and second output nodes,further comprising a capacitance coupled between said first and secondoutput nodes.
 4. A direct conversion receiver as in claim 1, where saiddown-conversion mixer operates as a differential signal down-conversionmixer having a pair of first input nodes, a pair of second input nodes,and first and second output nodes coupled to a power supply throughfirst and second load resistors, respectively, further comprising afirst capacitance coupled in parallel with said first load resistor anda second capacitance coupled in parallel with said second load resistor.5. A direct conversion receiver as in claim 4, further comprising athird capacitance coupled between said first and second output nodes. 6.A direct conversion receiver as in claim 1, where said down-conversionmixer operates as a differential signal down-conversion mixer having apair of first input nodes, a pair of second input nodes, and first andsecond output nodes coupled to a power supply through first and secondload resistors, respectively, further comprising a capacitance coupledbetween said first and second output nodes.
 7. A direct conversionreceiver as in claim 1, where said down-conversion mixer and said lowpass filter are implemented as part of an integrated circuit, where saidresistor and said capacitor are fabricated within said integratedcircuit.
 8. A direct conversion receiver as in claim 1, where an outputnode of said low pass filter is coupled to an input node of an automaticgain control circuit.
 9. A wireless communications mobile stationcomprising at least one antenna and a RE transceiver comprised of adirect conversion receiver coupled to said antenna, said directconversion receiver comprising a low noise amplifier for amplifying areceived RE signal and for outputting said amplified RF signal to adown-conversion current switching mixer comprising a first input nodefor receiving said amplified RE signal, a second input node forreceiving a local oscillator signal for mixing with said amplified REsignal and an output node coupled to an input of a low pass filter, saidoutput node further being coupled to a power supply voltage through aload resistance, said low pass filter having a low pass pole generatedby an amplifier having a resistor Rand a capacitor C coupled in parallelin a feedback path of said amplifier, where a low pass corner frequencyof said low pass filter is inversely proportional to the product of Rand C.
 10. A wireless communications mobile station as in claim 9,further comprising at least one capacitance coupled to said output nodeof said down-conversion mixer for attenuating higher order mixingfrequencies output from said down-conversion mixer.
 11. A wirelesscommunications mobile station as in claim 9, where said directconversion receiver comprises an in-phase (I) branch and a quadrature(Q) branch following said low noise amplifier, and where each of said Iand Q branches comprise one of said down-conversion mixers and one ofsaid low pass filters.
 12. A wireless communications mobile station asin claim 9, where said down-conversion mixer operates as a differentialsignal down-conversion mixer having a pair of first input nodes, a pairof second input nodes, and first and second output nodes coupled tofirst and second input nodes of said low pass filter, and where saidamplifier comprises an operational amplifier that has an RC pair coupledin parallel between an output node of said operational amplifier andsaid first input node of said low pass filter and has another RC paircoupled in parallel between said output node of said operationalamplifier and said second input node of said low pass filter.
 13. Amethod of operating a direct conversion receiver integrated circuit,comprising: amplifying an input RF signal; down-converting saidamplified RF signal to a down-converted signal using a current switchingmixer having an input transconductor stage for receiving said amplifiedRF signal, where the mixer processes the amplified RF signal in thecurrent mode and processes a local oscillator signal in the voltagemode; and low pass filtering said down-converted signal with an activelow pass filter having a low pass pole generated by an amplifier havinga resistor R and a capacitor C coupled in parallel in a feedback path ofsaid amplifier, where a low pass corner frequency of said low passfilter is inversely proportional to the product of R and C.
 14. A methodas in claim 13, where said current switching mixer operates as adifferential signal down-conversion mixer having a pair of first inputnodes, a pair of second input nodes, and first and second output nodescoupled to first and second input nodes of said active low pass filter,and where said amplifier comprises an operational amplifier that has anRC pair coupled in parallel between an output node of said operationalamplifier and said first input node of said low pass filter and hasanother RC pair coupled in parallel between said output node of saidoperational amplifier and said second input node of said low passfilter.
 15. A receiver circuit for use in a mobile station, comprising aGilbert Cell mixer that receives an RF signal to be mixed with a localoscillator signal, said Gilbert Cell mixer having first and secondoutputs connected to first and second inputs, respectively, of anoperational amplifier for providing differential currents i_(diff) tosaid first and second inputs of said operational amplifier, said firstand second inputs of said operational amplifier being connected to asupply voltage through first and second resistances, respectively, andreceiving common mode currents I_(DC) therefrom, said operationalamplifier having a first parallel RC network coupled between a firstoutput node of said operational amplifier and said first input and has asecond parallel RC network coupled between a second output node of saidoperational amplifier and said second input, said operational amplifierand said first and second parallel RC networks forming a low pass filterat said first and second outputs of said Gilbert Cell mixer.
 16. Areceiver circuit as in claim 15, where said first and second outputs ofsaid Gilbert Cell mixer are connected to virtual ground nodes of saidoperational amplifier.
 17. An integrated circuit for use with a radiofrequency receiver and comprising a down-conversion current switchingmixer comprising a first input node for coupling to a radio frequencysignal, a second input node for coupling to a local oscillator signalfor mixing with the radio frequency signal and an output node forcoupling to an input of an amplifier that comprises a portion of a lowpass filter having a low pass pole generated by a resistance R and acapacitance C coupled in parallel in a feedback path of said amplifier,said output node further for being coupled to a power supply voltagethrough a load resistance.
 18. An integrated circuit as in claim 17,where said down-conversion mixer operates as a differential signaldown-conversion mixer having a pair of first input nodes, a pair ofsecond input nodes, and first and second output nodes, furthercomprising a capacitance coupled between said first output nodes andsaid second output node.
 19. An integrated circuit as in claim 17, wheresaid down-conversion mixer operates as a differential signaldown-conversion mixer having a pair of first input nodes, a pair ofsecond input nodes, and first and second output nodes for coupling to apower supply through first and second load resistors, respectively,further comprising a first capacitance coupled in parallel with saidfirst load resistor and a second capacitance coupled in parallel withsaid second load resistor.
 20. An integrated circuit as in claim 17,where said resistance and said capacitance are fabricated within saidintegrated circuit.